XR76121

Voltage mode PWM

Voltage mode PWM is a simple technique that uses a single loop to control the output voltage. As shown in Figure 1, the output voltage is compared to a reference voltage with an error amplifier. The output of the error amplifier is then compared to a sawtooth and that output is used to drive the MOSFET, usually via a voltage divider.

Figure 1

As shown in Figure 2, the output voltage is modulated by turning the high-side FET on (on-time) with the pulse width and turning the low side FET off. At the end of the pulse, the high-side FET is turned off (off-time) and the low side FET is turned on until the next pulse. Vout = On-Time/Period * Vin.

The advantages of voltage mode PWM is that it is a very simple, common, smaller solution with good accuracy. The disadvantages are that complex frequency compensation is required (two poles) to stabilize the loop and because trailing edge control is most commonly used, there is a delay in load step response.

Current mode PWM

With voltage mode PWM, current is less known. For better control, current mode PWM senses the inductor current and it is compared to the reference voltage as shown in Figure 3.

Figure 3

Although the current has to be sensed with accuracy and introduces noise, the advantages of current mode PWM are easier loop compensation (less compensation needed with one pole), and it is easier to implement over-current protection and parallel currents to the output.

Standard Constant On-Time (COT)

As opposed to PWM, the pulse width in COT is always the same as shown in Figure 4. Instead the off-time length varies (as does the frequency) which modulates the output. As the Vout increases, the off-time of the duty cycle increases (frequency decreases) and the fixed on-time produces a lower duty cycle. This transfers less energy to the output and lowers the Vout. More simply said, as Vout increases, the duty cycle decreases. Conversely, as the Vout decreases, the off-time of the duty cycle decreases (frequency increases) and the fixed on-time produces a higher duty cycle. This transfers more energy to the output and raises the Vout.

Figure 4

The advantages of standard COT are very fast transient response, simplicity (inexpensive) and that frequency compensation is not complex as it is in PWM control. However, the feedback signal tends to have low amplitude and signal to noise ratio, making it very noise sensitive. Also, the output voltage is higher than the reference voltage and the ripple is dependent on and sensitive to the output capacitor ESR. This introduces a DC offset which is the average amount the output voltage is over the reference voltage. It is also jitter prone and the frequency changes during the load steps.

Some solutions solve the noise sensitivity by having one of two options that condition the feedback signal but introduce delays. One tradeoff provides faster transient responses; the other allows low ESR output capacitors to be used.

MaxLinear’s patented COT

MaxLinear’s patented COT architecture however conditions the reference instead as shown in Figure 5. The MaxLinear devices create their own emulated ramp that is insensitive to noise and the ESR of the capacitor. Since the output capacitor ESR does not affect it, low ESR ceramic capacitors can be used and maintain stability without decreasing speed. In addition, the Vout and reference voltage are compared and that result controls the ramp circuit. This creates a slower loop where the output voltage is averaged out and the DC offset is not introduced as in standard COT.

Figure 5

MaxLinear’s COT still has the standard COT advantages of very fast transient response, simplicity and no complex frequency compensation in addition to not having DC offset or ESR value sensitivity. MaxLinear’s COT architecture provides exceptional line and load regulation.

The XR76117 and XR76121 are identical to each other except for their output current rating. Similarly the IR3824/IR3825/IR3829 are identical to each other except for the differences shown in this section. Output current ratings are summarized in Table 1.

Table 1: Major Specification Comparison 

Table 2: Pin-out Comparison & Changes Required to Drop XR76117 / XR76121 into IR3824 / IR3825 / IR3829 Socket 

The PCB board can be easily designed to drop-in the XR76117 or XR76121 while maintaining compatibility to the IR3824/25/29. In Figure 2 below, pinning for both series are represented. The Exar XR76117 and XR76121 pin names do not have parenthesis, and the same corresponding pins for the IR3824/25/29 are in parenthesis. As shown in the legend, the components in green boxes are added and the ones in red boxes are omitted when using the XR76117 or XR76121. Both a pull-up and pull-down are shown for pins 2 and 6, but only one or the other will be present depending on the application. So, 4 passives (red) will not be populated while 3 passives and a jumper will be populated.

 

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In PWM controllers, frequency is constant and t_ON (on-time) is set by the controller to regulate the output voltage. However in COT controllers, the t_ON is constant and set by the R_ON resistor. This also sets the frequency. R_ON is connected between the TON pin and GND.

If the high-side FET was ideal, the t_ON of the SW signal would be equal to t_ON of GH (high-side FET gate) signal and the relationship between R_ON and t_ON would be:

 

However, the high side FET has rise and fall times as well as on and off delay times, so the t_ON of SW is not equal to GH. This non-ideal characteristic is measured for each regulator or module where the above equation is modified. In the Applications Information of each regulator or module datasheet, an equation defining the relationship of R_ON and t_ON is given based on test data for that device. For example, in the XR79206 power module datasheet, you will find the following equation in the Programming the On-Time section of the Applications Information:

 

 

 

 

 

 

 

 

The COT families (XRP6141, XRP6124 and XR75100 controllers, XR76xxx regulators and XR79xxx power modules) have 2 modes of operation that can be set: DCM / CCM (discontinuous conduction mode / continuous conduction mode) or FCCM (Forced CCM) mode. In FCCM mode, the converter operates at a preset frequency regardless of output current. In DCM / CCM mode the converter operates in DCM or CCM depending on the Iout magnitude. If Iout < ½ Ipp, the converter transitions to DCM mode. If Iout is higher, operation is in CCM mode.

The main advantage of DCM / CCM is that it provides significantly higher efficiency at light loads. For those applications where that doesn’t matter, FCCM can be used and has the advantage that it allows for operation at a constant frequency, regardless of load. It also results in lower Vout ripple, and will operate in an inaudible range.

There are many factors to consider in selecting the inductor including core material, inductance versus frequency, current handling capability, efficiency, size and EMI. Typically, the inductor is primarily chosen for value, saturation current and DC resistance (DCR). Increasing the inductor value will decrease output voltage ripple, but degrade transient response. Low inductor values provide the smallest size, but cause large ripple currents, poor efficiency and require more output capacitance to smooth out the larger ripple current. The inductor must be able to handle the peak current at the switching frequency without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. A good compromise between size, loss and cost is to set the inductor ripple current to be within 20% to 40% of the maximum output current.

The switching frequency and the inductor operating point determine the inductor value as follows:

L = Vout x (Vinmax – Vout) / Vinmax x fs x Kr x Ioutmax

Where fs = switching frequency

Kr = ratio of the AC inductor ripple current to the maximum output current

So for example, we want to choose L for the XR76108 (Ioutmax 8A) and wish to convert 12Vin to 2.5Vout with a frequency of 1MHz:

In our example, Ipp = 35% x 8A = 2.8A

Once the required inductor value is selected, the proper selection of core material is based on peak inductor current and efficiency requirements. The core must be large enough not to saturate at the peak inductor current.

Ipeak = Ioutmax + Ipp/2

In our example, Ipeak = 8A + 2.8A/2 = 9.4A

In general, it is set for Imax x 1.5. It would be close the maximum Iout (including ripple). If conservatively set too high, the hiccup mode may not be activated fast enough. If set too low, the ripple could cause the current to go over the threshold and set it into hiccup on a pre-mature basis.

A zero-cross comparator monitors the voltage across the low-side FET when it is on. The comparator threshold is nominally set at -1mV or -2mV (see individual datasheet). If there is sufficient I_OUT such that V_SW is below the threshold and therefore does not trigger the zero-cross comparator, CCM operation continues.

As I_OUT is reduced, V_SW gets closer to ground. When V_SW meets the threshold, the zero-cross comparator triggers. If there are 8 consecutive triggers, then DCM operation begins. The low side FET is turned off when I_L x R_DS equals the zero-cross threshold.

As there is no negative inductor current, the charge transferred to C_OUT is preserved. As I_OUT decreases further, less charge transfer to C_OUT is required. Pulses grow further apart, frequency is reduced and efficiency increases.

DCM persists as long as there are 8 consecutive zero-crosses.

Note that when the DCM frequency falls below about 1kHz, the controller turns on the lower-side FET for 100ns once every 1.2ms to refresh the charge on the bootstrap capacitor. This refresh cycle generates small spikes on SW, which can be seen interlaced between DCM pulses.